Is there a way to store and compute non-integer values in verilog, (say x = 5/2 = 2.5 ).
Can I compute and store 2.5 in x defined above?
Yes, you can use real registers to store real values, i.e.:
real float; // a register to store real value
Usually it's a 64-bit wide data type that stores floating-point values. But not all Verilog operators can be used with expression involving real numbers and real registers. Concatenations, modulus operator, case equality, bit-wise operators, reduction operators shift operators, bit-selects and part-selects on real type variables are not allowed.
Simple example:
module ecample;
real r;
initial begin
r = 123456e-3;
$display("r=%f",r); // r = 123.456000
#20 r = r / 2;
$display("r=%f",r); // r = 61.728000
$finish;
end
endmodule
Related
I have been working on approximate multiplication recently and I want to write a Verilog code for dynamic segment multiplication (DSM) . It suggest that you find the first index in you number which has a value of 1 and then take other 3 indexes next to it to form a 4 bit number that represent an 8 bit number then you should multiply these 4 bit numbers instead of 8 bits then some shifts to have the final result it helps a lot on hardware actually.. but my problem is about multiplication of these segments because sometimes they should be considered signed and some time unsigned I have the last 3 lines of my code: (a and b are input 8 bit numbers) and m1 and m2 are segments I wrote m,m2 as reg signed [3:0] and a and b as input signed [7:0]
Here is my code:
assign out = ({a[7],b[7]}==2'b11)||({a[7],b[7]}==2'b00) ? ($unsigned(m1)*$unsigned(m2)) << (shift_m1+shift_m2) : 16'dz;
assign out = ({a[7],b[7]}==2'b01) ? ($signed({1'b0,m1})*$signed(m2)) << (shift_m1+shift_m2) : 16'dz;
assign out = ({a[7],b[7]}==2'b10) ? ($signed(m1)*$signed({1'b0,m2})) << (shift_m1+shift_m2) : 16'dz;
But in simulation Verilog always considers segments as unsigned and does unsigned multiplication even though I noted signed or unsigned mark...
Can anyone help? I read all of the questions about this problem in stackoverflow and other places but still cannot solve this issue...
The rules for non-self determined operands say that if one operand is unsigned, the result is unsigned. 16'dz is unsigned.
The conditional operator i ? j : k has the condition operand i self-determined, but the two selections j and k are in a context based on the assignment or expression it is a part of. The shift operator i << j has the shift amount operand j self-determined.
All of the context rules are explained in section 11.6.1 Rules for expression bit lengths in the IEEE 1800-2017 SystemVerilog LRM.
You can get your desired result by using the signed literal 16'sdz.
However the logic you wrote may not be synthesizable for certain technologies that do not allow using a z state inside your device. The correct and more readable way is using a case statement:
alway #(*) case({a[7],b[7]})
2'b00,
2'b11: out = $unsigned(m1)*$unsigned(m2) << shift_m1+shift_m2;
2'b01: out = $signed({1'b0,m1})*m2 << shift_m1+shift_m2;
2'b10: out = m1*$signed({1'b0,m2}) << shift_m1+shift_m2;
endcase
I am somewhat new to verilog and I have a question that is confusing me .
I have a number of constant parameters , specifically nearly 1023 of them c0 , c1,c2 ..... c1022, each one being 10 bit in length . I also have a vector r[1022:0] , which is 1023 bits in length . My task is to compute ci*r[i] where i varies from 0 to 1022 and finally take the xor of the 1023 10 bit vectors that i get.When I do this in simulation , verilog generates the output at time 0 for the assign statement . How can verilog generate the output at time 0 ? Will there be no delay associated with these 1023 xors?
Also, if I need to do this succinctly , is there a short form that I can use or do I need to manually write c0 *r[0] ^ c1 *r[1] ......^ c[1022]*r[1022] which is synthesizable ?
A Verilog simulator will execute whatever legal syntax you give it—the tool knows nothing about what the implementation eventually looks like. It's up to you to feed timing constraints to the synthesis tool and it tells you if it can fit the logic to meet the constraints (or you might have to run another tool to see if it meets timing constraints).
Since you named your parameters c0, c1, c2, ..., you might as well named them czero, cone, ctwo, ... which gives you no options for shortcuts.
If you tool supports SystemVerilog, you can write your parameter as an array and then use the array xor reduction operator
parameter [9:0] C[1023] = {10'h123, 10'h234, ...};
assign out = C.xor() with (item*r[item.index]);
If you synthesis tool does not support this SystemVerilog syntax you, you can pack the parameter values into a single vector and use an indexed part select in Verilog.
parameter [10220-1:0] C = {10'h123, 10'h234, ...};
function [9:0] xor_reduction (input [1022:0] r);
integer I;
begin
xor_reduction = 0;
for(I=0;I<1023;I=I+1)
xor_reduction = xor_refuction ^ (r[1022-I]*C[I-:10]);
end
endfunction
assign out = xor_reduction(r);
I am trying to implement a linear feedback shift register with parameterized width and coefficients:
// ...
parameter width = 16;
parameter [width-1:0] coeff = 16'b1110101100110110;
// ...
Is there a way to assign the XOR-chain to the input flipflop, i.e. what is a sensible way of implementing something like
assign input_wire = (coeff[0] & flops[0]) xor ... xor (coeff[width-1] & flops[width-1]);
The obvious but illegal way would be using a for-loop. Is there a way to do this outside an always-block?
Thanks
Bitwise AND and the unary XOR-operator do the trick:
assign input_wire = ^(coefficients[width-1:0] & flops[width-1:0]);
The bitwise AND does the weighting of the flipflop outputs.
I'm somewhat stumped on this problem:
"Write a verilog module for full addition of n-bit integers. Let the parameter, the number of bits, equal 3. Call this module from a test bench, and in the test bench specify the numbers to be added in the arrays. Assign octal values to the X and Y arrays. The carryin is 0."
And yes, this is homework.
I was able to write the module for the n-bit adder:
module addern(carryin, X, Y, S, carryout, overflow);
parameter n = 3;
input carryin;
input [n-1:0] X, Y;
output reg [n-1:0] S;
output reg carryout, overflow;
always #(X,Y, carryin)
begin
{carryout, S} = X + Y + carryin;
overflow = (X[n-1] & Y[n-1] & ~S[n-1]) | (~X[n-1] & ~Y[n-1] & S[n-1]);
end
endmodule
I understand this component of the problem. However, I'm not sure how to implement the octal number addition. Is there a way in verilog to indicate that the arrays are holding octal values, rather than binary?
Is there anything like a typecast in verilog? For instance, input (octal) [n-1:0] X, Y, and do something likewise in the test bench.
Any constructive input is appreciated.
I'm pretty sure I'm in the same class as you. I think what you need to do is create a hierarchical Verilog module and then assign your values there. That would be your testbench. for example if you want to make X you write input [n-1:0] X = 3'o013, or maybe it's X = 9'o013 if Oli is correct. you don't change n, but it's kind of like BCD where they are in groups and you have a certain amount of bits you can represent before it overflows.
To help solve the problem thik about the question:
Q) How are numbers stored in digital hardware?
A) Binary, in digital logic we can only represent 2 values 1 and 0, but with this we can represent Integer, fixed point or floating point numbers.
Therefore digital numbers are base 2 (two possible values), while being able to represent any number. Other bases such as Octal (base 8) hex (base 16) and decimal (base 10) exist but these are just way of representing numbers, similar to the way binary just represents a number.
A decimal 1, is represented by 1 n all the bases, and when stored as binary they are all the same. An example of some values in verilog and there binary equivalents.
Octal Decimal Hex Binary
3'O7 => 3'd7 => 3'h7 => 3'b111
6'O10 => 6'd8 => 6'h8 => 6'b001000
Octal, Decimal and Hex in verilog are just representations of a binary format, a way of viewing the data. Since the low level electronics has no way of representing any thing other than 0 and 1.
The interesting thing about Octal and Hex is that they have a power of 2 values so they use an exact number of bits so an 9'O123 is the same as treating each Octal place separately and concatenating them together, 9'O123 == {3'O1, 3'O2, 3'O3}. This is also true for hexadecimal values but not decimal (base 10) values, as 10 is not a power of 2 and does not fully occupy the number space.
This does allow 'Octal' ports to be created, which are just 3 bit binary ports:
module octal_concat (
input [2:0] octal_2,
input [2:0] octal_1,
input [2:0] octal_0,
output [8:0] concat
);
assign concat = {octal_2, octal_1, octal_0};
endmodule
octal_concat octal_concat_0 (
.octal_2(3'O1),
.octal_1(3'O2),
.octal_0(3'O3),
.concat() //Drives 9'O123 which is also 9'b001_010_011
);
This is a follow-on question from How can I iteratively create buses of parameterized size to connect modules also iteratively created?. The answer is too complex to answer in a comment and the solution may be helpful for other SOs. This question is following the self-answer format. Addition answer are encouraged.
The following code works and uses a bi-directional array.
module Multiplier #(parameter M = 4, parameter N = 4)(
input [M-1:0] A, //Input A, size M
input [N-1:0] B, //Input B, size N
output [M+N-1:0] P ); //Output P (product), size M+N
wire [M+N-1:0] PP [N-1:0]; // Partial Product array
assign PP[0] = { {N{1'b0}} , { A & {M{B[0]}} } }; // Pad upper bits with 0s
assign P = PP[N-1]; // Product
genvar i;
generate
for (i=1; i < N; i=i+1)
begin: addPartialProduct
wire [M+i-1:0] gA,gB,gS; wire Cout;
assign gA = { A & {M{B[i]}} , {i{1'b0}} };
assign gB = PP[i-1][M+i-1:0];
assign PP[i] = { {(N-i){1'b0}}, Cout, gS}; // Pad upper bits with 0s
RippleCarryAdder#(M+i) adder( .A(gA), .B(gB), .S(gS), .Cin(1'b0), .* );
end
endgenerate
endmodule
Some of the bits are never used, such as PP[0][M+N-1:M+1]. A synthesizer will usually remove these bits during optimization and possibly give a warning. Some synthesizers are not advance enough to do this correctly. To resolve this, the designer must implement extra logic. In this example the parameter for all the RippleCarryAdder's would be set to M+N. The extra logic wastes area and potently degrades performance.
How can the unused bits be safely eliminated? Can multidimensional arrays with different dimensions be used? Will the end code be readable and debug-able?
Can multidimensional arrays with different dimensions be used?
Short answer, NO.
Verilog does not support unique sized multidimensional arrays. SystemVerilog does support dynamic arrays however these cannot be connected to module ports and cannot be synthesized.
Embedded code (such as Perl's EP3, Ruby's eRuby/ruby_it, Python's prepro, etc.) can generate custom denominational arrays and code iterations, but the parameters must be hard coded before compile. The final value of any parameter of a given instance is discoverer during compile time, well after the embedded script is ran. The parameter must be treated as a global constant, therefore Multiplier#(4,4) and Multiplier#(8,8) cannot exist in the same project unless to teach the script how to extract the full hierarchy and parameters of the project. (Good luck coding and maintaining that).
How can the unused bits be safely eliminated?
If the synthesizer is not advance enough to exclude unused bits on its own, then the bits can be optimized by flattening the multidimensional array into a one-dimensional array with intelligent part-select. The trick is finding the equation which can be achieved by following these steps:
Find the pattern of the lsb index for each part part select:
Assume M is 4, the lsb for each part-select are 0, 5, 11, 18, 26, 35, .... Plug this pattern into WolframAlpha to find the equation a(n) = (n-1)*(n+8)/2.
Repeat with M equal to 3 for the pattern 0, 4, 9, 15, ... to get equation a(n)=(n-1)*(n+6)/2
Repeat again with M equal to 5 for the pattern 0, 6, 13, 21, 30, ... to get equation a(n)=(n-1)*(n+10)/2.
Since the relation of M and N is linear (i.e. multiple; no exponential, logarithmic, etc.), only two equations are needed to create a variable parameter M equation. For non-linear equations more data-point equations are recommended. In this case note that for M=3,4,5 the pattern (n+6),(n+8),(n+10), therefore the generic equation can be derived to: lsb(n)=(n-1)*(n+2*M)/2
Fine the pattern of the msb index for each part select:
Use the same process of as finding the lsb (ends up being msb(n)=(n**2+(M*2+1)*n-2)/2). Or define the msb in terms of lsb: msb(n)=lsb(n+1)-1
IEEE std 1364-2001 (Verilog 2001) introduced macros with arguments and indexed part-select; see § 19.3.1 '`define' and § 4.2.1 'Vector bit-select and part-select addressing' respectively. Or see IEEE std 1800-2012 § 22.5.1 '`define' and § 11.5.1 'Vector bit-select and part-select addressing' respectively. This answer assumes that these features are supported by the SO's simulator and synthesizer since the generate keyword was also introduced in IEEE std 1364-2001, see § 12.1.3 'Generated instantiation' (and IEEE std 1800-2012 § 27. 'Generate constructs'). For tools that are not fully support IEEE std 1364-2001, see `ifdef examples provided here.
Since the functions to calculate the part-select ranges are frequently used, use `define macros with arguments. This will help prevent copy/paste bugs. The extra sets of () in the macro definitions are to insure proper order of operations. It is also a good idea to `undef the macros at the end of the module definition, preventing the global space from getting polluted. With the flattened array it may become challenging to debug. By defining pass-through connections within the generate block's for-loop the signal can become readable and can be probed in waveform.
module Multiplier #(parameter M = 4, parameter N = 4)(
input [M-1:0] A, //Input A, size M
input [N-1:0] B, //Input B, size N
output [M+N-1:0] P ); //Output P (product), size M+N
// global space macros
`define calc_pp_lsb(n) (((n)-1)*((n)+2*M)/2)
`define calc_pp_msb(n) (`calc_pp_lsb(n+1)-1)
`define calc_pp_range(n) `calc_pp_lsb(n) +: (M+n)
wire [`calc_pp_msb(N):0] PP; // Partial Product
assign PP[`calc_pp_range(1)] = { 1'b0 , { A & {M{B[0]}} } };
assign P = PP[`calc_pp_range(N)]; // Product
genvar i;
generate
for (i=1; i < N; i=i+1)
begin: addPartialProduct
wire [M+i-1:0] gA,gB,gS; wire Cout;
assign gA = PP[`calc_pp_range(i)];
assign gB = { A & {M{B[i]}} , {i{1'b0}} };
assign PP[`calc_pp_range(i+1)] = {Cout,gS};
RippleCarryAdder#(M+i) adder( .A(gA), .B(gB), .S(gS), .Cin (1'b0), .* );
end
endgenerate
// Cleanup global space
`undef calc_pp_range
`undef calc_pp_msb
`undef calc_pp_lsb
endmodule
Working example with side-by-side and test bench: http://www.edaplayground.com/s/6/591
Will the end code be readable and debug-able?
Yes, for anyone who has already learned how to properly use the generate construct. The generate block's for-loop defines local wires which are confined to scope of the loop index. gA form loop-0 and gA from loop-1 are unique signals and cannot interact with each other. The local signals can be probed in waveform which is great for debugging.